TRANSFORMER CIRCUIT MAGNETIZING REACTANCE

For an ideal transformer, the magnetizing current is assumed to be negligible. For a real transformer, some magnetizing current must flow when voltage is applied to the winding in order to establish a flux in the core.

The voltage induced in the winding by the flux restrains the magnetizing current. It was shown earlier that the magnetizing current is not really sinusoidal, but contains many odd harmonics in addition to the fundamental frequency.

If we neglect the harmonics and concentrate on the fundamental frequency, the magnetizing current in the winding lags the applied voltage by 90°. In a two-winding transformer, this is equivalent to placing a reactance Xm, called the magnetizing reactance, in parallel with the transformer terminals.

The peak value of the magnetizing current is determined from the B-H curve of the core, which we have seen is very nonlinear. Therefore, the magnetizing reactance is not a constant but is voltage dependent; however, if the peak flux density is kept well below the point of saturation, Xm can be approximated by a constant reactance in most engineering calculations.

It is generally desirable to maximize Xm in order to minimize the magnetizing current. We saw earlier that inductance is inversely proportional to the reluctance of the core along the flux path and the reluctance of an air gap is several thousand times the reluctance of the same distance through the steel.

Therefore, even tiny air gaps in the flux path can drastically increase the core’s reluctance and decrease Xm. A proper core design must therefore eliminate all air gaps in the flux path.

Since any flux that is diverted must flow between the laminations through their surfaces, it is vital that these surfaces lie perfectly flat against each other. All ripples or waves must be eliminated by compressing the core laminations together tightly.

This also points out why the oxide layers on the lamination surfaces must be extremely thin: since these layers have essentially the same permeability as air and since the flux that is diverted from the air gaps must then travel through these oxide layers, the core’s reluctance would greatly increase if these layers were not kept extremely thin.

TRANSFORMER NO LOAD LOSSES BASICS AND TUTORIALS

TRANSFORMER NO LOAD LOSSES BASIC INFORMATION
What Are The Transformer No Load Losses?


Alternating magnetic flux produces both hysteresis losses and eddy-current losses in the steel. As we have seen, hysteresis losses depend on several factors including the frequency, the peak flux density, the type of core steel used, and the orientation of the flux with respect to the ‘‘grain’’ of the steel.

All of the above factors, except the frequency, are under the control of the transformer designer. Core losses are sometimes referred to as iron losses and are commonly referred to as no load losses, because core losses do not require load currents.

Decreasing the induced voltage per turn can reduce the peak flux density. This obviously involves increasing the numbers of turns in both the primary and secondary windings in order to maintain the same transformer turns ratio.

The disadvantage of adding more turns is that this increases the length of conductor and increases the conductor resistance. More cross sectional area is required in order to keep the resistance constant.

Doubling the number of turns requires about four times the volume of copper. Another way of reducing core losses is to use various types of low-loss core steels that are now available, including ‘‘amorphous’’ core materials, which have extremely low losses and superior magnetic properties.

Unfortunately, amorphous core materials have ceramic-like properties, so fabricating transformer cores with these materials is much more difficult than with laminated steel cores.

With grain-oriented steel, the direction of the core flux must be kept more or less parallel to the grain of the steel by mitering the corners of the laminations where the flux changes direction by 90°. Since the flux will cross the grain at about a 45° angle at the mitered edges, the hysteresis losses will increase somewhat in these places.

These additional localized core losses must be factored into the calculation of the total core losses. Building up the core with thin laminated strips controls eddy losses in the core, each strip having an oxide film applied to the surface.

The oxide film is extremely thin and it is more like a high-resistance film than true electrical insulation; but since the potential differences between adjacent laminations is quite small, the resistance of the oxide film is very effective in breaking up the eddy current paths.

During the manufacture of the core, the core cutting machine must not be allowed to get dull; otherwise, ‘‘burrs’’ will form along the edges of the laminations. Burrs are imperfections that form electrical bridges between the laminations and create paths for eddy currents and increased losses.

Sometimes the eddy currents near a burr can be large enough to cause localized overheating that can actually cause core damage. Core losses are approximately proportional to the square of the excitation voltage E applied to the transformer.

Therefore, placing an equivalent linear conductance Gm across the transformer terminals can approximate transformer core losses. The core losses are expressed by Wm = E^2Gm